Chapter 4. Voltage amplifiers and controls

141



Voltage Amplifiers and Controls

Base .-,~



[



..Emitter

'NPN'



.__1< c

E



Mountingsubstrate



Base- ~



)



_f,

,,P~,



P



co~,~,o~



, Emitter



~,



,1"~



///,~/ //,;,'////////////

Mountingsubstrate



Fig. 4.1



c



'PNP'



~_



E



//



Circuitsymbols



Typical chip cross-section of NPN and PNP silicon planar

epitaxial transistors.



the major problem in obtaining good linearity lies in the nature of the

base voltage/collector current transfer characteristic, shown in the case of

a typical 'NPN' device (a 'PNP' device would have a very similar characteristic, but with negative voltages and currents) in Fig. 4.2.



+ mA



Collector



current

(Ia



+



I



0V



Fig. 4.2



x



(O.S3 V)



y



(0-7 V)



--



Basevoltage (Vb)



Typical transfer characteristic of sificon transistor.



142



Voltage Amplifiers and Controls



.J



/I A



r



k/



!



k../



Ol~ptd C'Uff~nd

swing



+V~



f



I



Fig. 4.3



Inputvoltage

sww~



J



Transistor amplifier waveform distortion due to transfer

characteristics.



In this, it can be seen that the input/output transfer characteristic is

strongly curved in the region 'X - Y' and an input signal applied to the

base of such a device, which is forward biased to operate within this

region, would suffer from the very prominent (second harmonic) waveform

distortion shown in Fig. 4.3.

The way this type of non-linearity is influenced by the signal output

level is shown in Fig. 4.4. It is normally found that the distortion

increases as the output signal increases, and conversely.

There are two major improvements in the performance of such a

bipolar amplifier stage which can be envisaged from these characteristics.

Firstly, since the non-linearity is due to the curvature of the input characteristics of the device - the output characteristics, shown in Fig. 4.5, are

l i n e a r - the smaller the input signal which is applied to such a stage, the

lower the non-linearity, so that a higher stage gain will lead to reduced

signal distortion at the same output level. Secondly, the distortion due to

such a stage is very largely second harmonic in nature.



143



Voltage Amplifiers and Controls



Output ~ e l

r



0



Fig. 4.4



Relationship between signal distortion and output signal voltage

in bipolar transistor amplifier.

|r

(mA)



~, V~=0.7 V



9-



Va-0.65 V



V~=O.eV

.



.



.



.



Vb-0"S5 V

, ,



.



V~-O.SV

Collector voltage

(Vc)



Fig. 4.5



Output current~voltage characteristics of typical silicon bipolar

transistor.



This implies that a 'push-pull' arrangement, such as the so-called 'longtailed pair' circuit shown in Fig. 4.6, which tends to cancel second harmonic distortion components, will greatly improve the distortion

characteristics of such a stage.

Also, since the output voltage swing for a given input signal (the stage

gain) will increase as the collector load (R2 in Fig. 4.6) increases, the

higher the effective impedance of this, the lower the distortion which will

be introduced by the stage, for any given output voltage signal.

If a high value resistor is used as the collector load for Q~ in Fig. 4.6,

either a very high supply line voltage must be applied, which may exceed

the voltage ratings of the devices or the collector current will be very



Voltage Amplifiers and Controls



144



+V



0,1



I

R4 (=R1)



OV



R,



---OV



-V



Fig. 4.6



Transistor voltage amplifier using long-tailed pair circuit layout.



small, which will reduce the gain of the device, and therefore tend to

diminish the benefit arising from the use of a higher value load resistor.

Various circuit techniques have been evolved to circumvent this problem,

by producing high dynamic impedance loads, which nevertheless permit

the amplifying device to operate at an optimum value of collector current.

These techniques will be discussed below.

An unavoidableproblem associated with the use of high values of

collector load impedance as a means of attaining high stage gains in such

amplifier stages is that the effect of the 'stray' capacitances, shown as C~

in Fig. 4.7, is to cause the stage gain to decrease at high frequencies as

the impedance of the stray capacitance decreases and progressively begins

to shunt the load. This effect is shown in Fig. 4.8, in which the 'transition'

frequency, fo, (the - 3 dB gain point) is that frequency at which the shunt

impedance of the stray capacitance is equal to that of the load resistor, or

its effective equivalent, if the circuit design is such that an 'active load' is

used in its place.



Field effect devices



Other devices which may be used as amplifying components are field

effect transistors and MOS devices. Both of these components are very

much more linear in their transfer characteristics but have a very much

lower mutual conductance (Gm).

This is a measure of the rate of change of output current as a function

of an applied change in input voltage. For all bipolar devices, this is

strongly dependent on collector current, and is, for a small signal silicon

transistor, typically of the order of 45 mA/V, per mA collector current.



145



Voltage Amplifiers and Controls



R2



o[



~,



-



. . . . . _



Eout



I

I

I



"" r "" C.



I



J.

~'//1////

Fig. 4.7



Circuit effect of stray capacitance.



dB I Outputsignalvoltage



fo

I

I



,



Fig. 4.8



h .



Influence of circuit stray capacitances on stage gain.



Power transistors, operating at relatively high collector currents, for which

a similar relationship applies, may therefore offer mutual conductances in

the range of amperes/volt.

Since the output jmpedance of an emitter follower is approximately

1/Gm, power output transistors used in this configuration can offer very

low values of output impedance, even without externally applied negative

feedback.

All field effect devices have very much lower values for Gin, which will

lie, for small-signal components, in the range 2 - 1 0 m A / V , not significantly

affected by drain currents. This means that amplifier stages employing

field effect transistors, though much more linear, offer much lower stage

gains, other things being equal.



Voltage Amplifiers and Controls



146



The transfer characteristics of junction (bipolar) FETs, and enhancement

and depletion mode MOSFETS are shown in Figs 4.9(a), (b) and (c).

MOSFETs



MOSFETs, in which the gate electrode is isolated from the source/drain

channel, have very similar transfer characteristics to that of junction

FETs. They have an advantage that, since the gate is isolated from the

drain/source channel by a layer of insulation, usually silicon oxide or

nitride, there is no maximum forward gate voltage which can be applied within the voltage breakdown limits of the insulating layer. In a junction

FET the gate, which is simply a reverse biassed PN diode junction, will

conduct if a forward voltage somewhat in excess of 0.6 V is applied.

The chip constructions and circuit symbols employed for small signal

lateral MOSFETs and junction FETs (known simply as FETs) are shown

in Figs 4.10 and 4.11.

It is often found that the chip construction employed for junction FETs

is symmetrical, so that the source and drain are interchangeable in use.



I



I



JL



_Vo



I



I



0



|1



0-6V



+Vg



0



+V o



' E n l ~ l c e n ~ t ' type MOSFET



Junclion FET



(b)



(a)



w



J



- Ve



t



0



+ Ve



'DepleUon' lype MOSFET



(c)



Fig. 4.9



Gate voltage versus drain current characteristics of field effect

devices.



Voltage Amplifiers and Controls

Gate



D



G q :ub



~etrate and mount

(Source connected to ,mbetrate)

Fig. 4 . 1 0



147



N-ch~



Chip cross-section and circuit symbol for lateral MOSFET

(small signal type)

Gate



P



(Gin oo,'mec',udto _ , . _ ~ )

D

G



(s)

O/S



G

S



~)



St)



Fig. 4.11 Chip cross-section and circuit symbols for (bipolar) junction



FET.

For such devices the circuit symbol shown in Fig. 4.11(c) should properly

be used.

A practical problem with lateral devices, in which the current flow

through the device is parallel to the surface of the chip, is that the path

length from source to drain, and hence the device impedance and current

carrying capacity, is limited by the practical problems of defining and

etching separate regions which are in close proximity, during the manufacture of the device.

V-MOS AND T'-MOS



This problem is not of very great importance for small signal devices, but

it is a major concern in high current ones such as those employed in

power output stages. It has led to the development of MOSFETs in which

the current flow is substantially in a direction which is vertical to the

surface, and in which the separation between layers is determined by

diffusion processes rather than by photo-lithographic means.



148



Voltage Amplifiers and Controls



Devices of this kind, known as V-MOS and T-MOS constructions, are

shown in Figs 4.12(a) and (b).

Although these were originally introduced for power output stages, the

electrical characteristics of such components are so good that these have

been introduced, in smaller power versions, specifically for use in small

signal linear amplifier stages. Their major advantages over bipolar devices,

having equivalent chip sizes and dissipation ratings, are their high input

impedance, their greater linearity, and their freedom from 'hole storage'

effects if driven into saturation.

These qualities are increasingly attracting the attention of circuit designers working in the audio field, where there is a trend towards the

design of amplifiers having a very high intrinsic linearity, rather than

relying on the use of negative feedback to linearise an otherwise worse

design.

BREAKDOWN



A specific problem which arises in small signal MOSFET devices is that,

because the gate-source capacitance is very small, it is possible to induce

breakdown of the insulating layer, which destroys the device, as a result

of transferred static electrical charges arising from mishandling.

Though widely publicised and the source of much apprehension, this

Source



GatemetafJbta~on



"*



~./ef



CurrentIk)w



"~



Drainand substrate

(a)

Gate

Source



Ox~e

layer



d ~ ~ ~ ~



Polysilicongate



,



|



,



N+

Drain 8nd ~ t e



(b)



Fig. 4.12 Power MOSFET constructions using (a) V and (b) T configurations. (Practical devices will employ many such cells in parallel.)



149



Voltage Amplifiers and Controls



problem is actually very rarely encountered in use, since small signal

MOSFETs usually incorporate protective zener diodes to prevent this

eventuality, and power MOSFETs, where such diodes may not be used

because they may lead to inadvertent 'thyristor' action, have such a high

gate-source capacitance that this problem does not normally arise.

In fact, when such power MOSFETs do fail, it is usually found to be

due to circuit design defects, which have either allowed excessive operating

potentials to be applied to the device, or have permitted inadvertent VHF

oscillation, which has led to thermal failure.



NOISE LEVELS

Improved manufacturing techniques have lessened the differences between

the various types of semiconductor device, in respect of intrinsic noise

level. For most practical purposes it can now be assumed that the

characteristics of the device will be defined by the thermal noise figure of

the circuit impedances. This relationship is shown in the graph of Fig.

4.13.

For very low noise systems, operating at circuit impedance levels which

have been deliberately chosen to be as low as practicable - such as in

moving coil pick-up head amplifiers - bipolar junction transistors are

still the preferred device. These will either be chosen to have a large base

junction area, or will be employed as a parallel-connected array; as, for

example, in the LM194/394 'super-match pair' ICs, where a multiplicity of

parallel connected transistors are fabricated on a single chip, giving an

effective input (noise) impedance as low as 40 ohms.

RMS noise vonage

3(X) - - ( n ~



200-



/



/



/



/



~/"



. .



/



whq~ K= 1"38x 10-"



T_=~



100-



tempmtum



re'tot GF bmxlvvtdUl



Cm~t impedance (n)

0 - .,,1



Fig. 4.13



, . . . .



lO



I



lOO



I



1K



'



'



t



10 K



Thermal noise output as a function of circuit impedance.



150



Voltage Amplifiers and Controls



However, recent designs of monolithic dual J-FETs, using a similar

type of multiple paraUel-connection system, such as the Hitachi 2SK389,

can offer equivalent thermal noise resistance values as low as 33 ohms,

and a superior overall noise figure at input resistance values in excess of

100 ohms.

At impedance levels beyond about 1 kilohm there is little practical

difference between any devices of recent design. Earlier MOSFET types

were not so satisfactory, due to excess noise effects arising from carrier

trapping mechanisms in impurities at the channel/gate interface.



OUTPUT VOLTAGE CHARACTERISTICS

Since it is desirable that output overload and signal clipping do not occur

in audio systems, particularly in stages preceding the gain controls, much

emphasis has been placed on the so-called 'headroom' of signal handling

stages, especially in hi-fi publications where the reviewers are able to

distance themselves from the practical problems of circuit design.

While it is obviously desirable that inadvertent overload shall not occur

in stages preceding signal level controls, high levels of feasible output

voltage swing demand the use of high voltage supply rails, and this, in

turn, demands the use of active components which can support such

working voltage levels.

Not only are such devices more costly, but they will usually have

poorer performance characteristics than similar devices of lower voltage

ratings. Also, the requirement for the use of high voltage operation may

preclude the use of components having valuable characteristics, but which

are restricted to lower voltage operation.

Practical audio circuit designs will therefore regard headroom simply as

one of a group of desirable parameters in a working system, whose design

will be based on careful consideration of the maximum input signal levels

likely to be found in practice.

Nevertheless, improved transistor or IC types, and new developments

in circuit architecture, are welcomed as they occur, and have eased the

task of the audio design engineer, for whom the advent of new programme

sources, in particular the compact disc, and now digital audio tape systems,

has greatly extended the likely dynamic range of the output signal.



Signal characteristics

The practical implications of this can be seen from a consideration of the

signal characteristics of existing programme sources. Of these, in the past,



Voltage Amplifiers and Controls



151



the standard vinyl ('black') disc has been the major determining factor. In

this, practical considerations of groove tracking have limited the recorded

needle tip velocity to about 40 cm/s, and typical high-quality pick-up

cartridges capable of tracking this recorded velocity will have a voltage

output of some 3 mV at a standard 5 cm/s recording level.

If the pre-amplifier specification calls for maximum output to be obtainable at a 5 cm/s input, then the design should be chosen so that there is a

'headroom factor' of at least 8 x , in such stages preceding the gain

controls.

I n general, neither FM broadcasts, where the dynamic range of the

transmitted signal is limited by the economics of transmitter power, nor

cassette recorders, where the dynamic range is constrained by the limited

tape overload characteristics, have offered such a high practicable dynamic

range.

It is undeniable that the analogue tape recorder, when used at 15 in.Is,

twin-track, will exceed the LP record in dynamic range. After all, such

recorders were originally used for mastering the discs. But such programme

sources are rarely found except among 'live recording' enthusiasts. However, the compact disc, which is becoming increasingly common among

purely domestic hi-fi systems, presents a new challenge, since the practicable dynamic range of this system exceeds 80 dB (10000:1), and the

likely range from mean (average listening level) to peak may well be as

high as 35 dB (56:1) in comparison with the 18 dB (8:1) range likely with

the vinyl disc.

Fortunately, since the output of the compact disc player is at a high

level, typically 2 V RMS, and requires no signal or frequency response

conditioning prior to use, the gain control can be sited directly at the

input of the preamp. Nevertheless, this still leaves the possibility that

signal peaks may occur during use which are some 56x greater than the

mean programme level, with the consequence of the following amplifier

stages being driven hard into overload.

This has refocused attention on the design of solid state voltage amplifier

stages having a high possible output voltage swing, and upon power

amplifiers which either have very high peak output power ratings, or

more graceful overload responses.



VOLTAGE AMPLIFIER DESIGN

The sources of non-linearity in bipolar junction transistors have already

been referred to, in respect of the influence of collector load impedance,

and push-pull symmetry in reducing harmonic distortion. An additional

factor with bipolar junction devices is the external impedance in the base

circuit, since the principal non-linearity in a bipolar device is that due to